1. Field of Invention
The present invention relates to a voltage reference circuit with 2nd order temperature compensation.
2. Description of Related Art
Reference circuits are necessarily present in many applications, such as purely analog, mixed-mode, to purely digital circuits. The demand for low voltage references is especially apparent in mobile battery operated products, such as cellular phones, pagers, camera recorders, and laptops. Consequently, low voltage and low quiescent current flow are required characteristics for improving battery efficiency and longevity. Low voltage operation is a consequence of improved process technology. Unfortunately, lower dynamic range (a consequence of low voltage) demands that reference voltages be more accurate.
Voltage references are generally required to provide a substantially constant output voltage despite gradual or momentary changes in input voltage, output current or temperature. In particular, many designers have utilized bandgap reference circuits due to their ability to provide a stable voltage supply that is insensitive to temperature variations over a wide temperature range. These bandgap references rely on certain temperature-dependent characteristics of the base-emitter voltage Vbe of a BJT transistor. Typically, these bandgap reference circuits operate on the principle of compensating the negative temperature coefficient of a bipolar transistor's base-emitter voltage with the positive temperature coefficient of the thermal voltage, i.e., with VThermal=kT/q, where k is Boltzmann's constant, T is the absolute temperature in degrees Kelvin, and q is the electronic charge. In general, the negative temperature coefficient of the base-emitter voltage is summed with the positive temperature coefficient of the thermal voltage VThermal, which is appropriately scaled such that the resultant summation provides a zero temperature coefficient.
While the bandgap reference is ideally desired to be independent of temperature, or at least linear with temperature, in practice the bandgap reference will typically produce a reference voltage only to be independent of temperature in a given range of temperature. This characteristic of the bandgap reference is mainly due to the fact that the Vbe (T) term is a non-linear function. In other words, an inherent variation exists for the base-emitter voltage Vbe of a transistor with respect to temperature. In particular, the bandgap reference generates a strong second-order term that varies with T ln(T), and which limits the temperature drift performance of such a reference, i.e., causes deviation of the reference voltage with temperature. While these second order terms may be relatively small, their impact can prove highly undesirable for many applications.
Various methods have been used to compensate for the temperature curvature characteristics for bandgap references. These methods have included the addition of circuitry which first attempts to measure the temperature curvature of the base-emitter voltage Vbe, and then sum the measured temperature curvature term with the bandgap reference output. Other methods have included the addition of circuitry that approximates the temperature curvature with a squared function of the temperature, such as by utilizing a proportional-to-absolute-temperature (PTAT) current through a resistor having a given temperature coefficient TC. While these methods may be utilized with some success, limitations exist over process availability and process variations. Most notably, many of these methods have been configured to address applications utilizing bipolar transistors, but can not be utilized effectively with CMOS applications. This limitation of prior art methods results from that the parasitic vertical bipolar transistor available in standard CMOS process has its collector always connected to substrate and limits the use of vertical bipolar transistor as an emitter follower.
FIG. 1 shows a conventional reference circuit having mixed current and voltage-mode architecture. FIG. 2 is a graph showing the temperature dependence of the reference circuit of FIG. 1.
As shown in FIG. 1, a bandgap circuit 16 utilizes a current-mode approach with a voltage-mode ladder. The bandgap circuit 16 includes series connected current source AIVbe and resistors R13, R12, and R11 coupled between a voltage source V and GND. Bandgap reference voltage Vref is produced at node between current source AIVbe and resistor R13. Current source BIPTAT is coupled between voltage source V and node a between resistors R13 and R12. Current source CINL is coupled between voltage source V and node b between resistors R12 and R11.
The resulting relation of the reference voltage Vref can be described by:Vref=AIVbe*(R11+R12+R13)+BIPTAT*(R11+R12)+CINL*R11
where IVbe, IPTAT, and INL correspond to the base-emitter, PTAT, and nonlinear temperature dependent currents respectively.
The curvature corrected bandgap of FIG. 1 has a temperature dependence as illustrated in FIG. 2. It achieved a temperature drift of 8.6 μV/° C. (−15° C. to 90° C.).
However, IVbe and IPTAT are not directly related. Besides, due to process variation, IVbe may be larger than expected. So, INL is smaller than expected (i.e. INL may be process dependent). In fact, when IVbe is larger than expected, Vref will suffer faster decay with increasing temperature and a larger INL is required to compensate for IVbe. However, actually, INL just moves in the opposite way and worsens offset of the bandgap reference voltage.
FIG. 3 shows another bandgap reference circuit 300 having amplifier 304, for example a dual differential amplifier. In addition, amplifier 304 is suitably configured such that one pair of differential inputs are suitably coupled to transistors 305 and 306 and resistors 301, 302 and 303. Further, the second pair of differential inputs of amplifier 304 can be suitably coupled to two transistors having different temperature coefficients, such as 306 having a PTAT/R current and 307 having a Vbe/R current. Accordingly, one pair of differential inputs can receive a voltage reference, while the second pair of differential inputs can receive a temperature curvature compensation voltage, i.e., one having a Tln(T) term. As a result of the feedback arrangement, any offset voltage realized by the second differential pair will be inverted and realized by the first differential pair, with or without suitable scaling by the effective gm contributions within dual differential amplifier 304, to provide a temperature compensated reference voltage Vout.
However, too many BJT transistors are used to implement the architecture of FIG. 3 and accordingly matching between BJT transistors is poor. In addition, with the same percentage of decrease in R301/R302/R303, increase in VE3 (emitter voltage of the transistor 307) is larger than that in VE2 (emitter voltage of the transistor 306). So, curvature compensation effect provided by the architecture of FIG. 3 is process sensitive, which is undesirable. Further, as for the dual differential pairs in the OP 304, 4 PMOS transistors have to be matched well, which is difficult because each pair has its own N-well.
Therefore, a high precision and trim-free bandgap circuit with process independent curvature compensation scheme is preferred.